3G radio

ABSTRACT

An automatic gain control is provided for a multi-mode receiver. The gain control includes feed forward controller  402  which sets up an initial gain rapidly based on a measurement of signal strength or upon mode information, and a feedback controller  450  which then takes responsibility for maintaining a signal at a desired amplitude.

[0001] The present invention relates to device architecture of mobiletelephony units.

[0002] There are an emerging number of standards for cellularcommunication. For example, the European GSM system works intransmission bands known as GSM 850, GSM 900, GSM 1800 and GSM 1900,where the numeric part of the name is indicative of the frequency of theband expressed in MHz. Furthermore, the UMTS system operates on atransmission band between 1.92 and 1.98 GHz. It would clearly bedesirable if a telecommunications device could easily switch betweenthese various telecommunications standards depending on which service itwished to use, or indeed which service was available.

[0003] According to a first aspect of the present invention, there isprovided a transmitter for GSM and UMTS comprising: anin-phase/quadrature up-converter for mixing in-phase and quadratureinputs with an intermediate frequency; a GSM path including a phaselocked loop; and a UMTS path;

[0004] wherein a frequency generator module is provided to generate afirst signal at a frequency F₁, and the first signal is supplied as aninput to an image reject mixer in the GSM path, to a mixer in the UMTSpath, and as an input to divider which divides the first signal by threeto create an intermediate frequency which is supplied to thein-phase/quadrature up converter; and wherein the image reject mixer inthe GSM path is controllable to select either an upper or lower sideband such that the GSM path operates at either (1+⅓)F₁ or (1−⅓)F₁; andwherein the mixer in the UMTS path selects the upper side band so as tohave a output at (1+⅓)F₁.

[0005] It is thus possible to provide a transmitter arrangement operablein both the GSM bands, and the UMTS band in which many of the RFcomponents are shared. Thus, for example, if the first signal from thefrequency generator is in a band centered about 1.35 GHz, but extendingas low as 1.28 GHz and as high as 1.485 GHz then it is possible to tunethe transmitter to selectively operate in the GSM 850 and GSM 900 bands,and in the GSM 1800 and 1900 bands and in the UMTS band between 1.92 and1.98 GHz.

[0006] Advantageously the first radio frequency signal is in factgenerated by a ultra high frequency voltage controlled oscillatorworking in the range of 2.565 to 2.97 GHz. This frequency can then bedivided by 2 by a divider in order to ensure that the first RF frequencyhas an equal mark space ratio. It will be appreciated by the personskilled in the art that transistor switching and logic technologies arenow fast enough to operate at these frequencies. Furthermore, thedivider can be arranged to produce in-phase and quadrature versions ofthe first radio frequency signal.

[0007] Advantageously the divider for generating the intermediatefrequency (i.e., a local oscillator signal) is a regenerative dividerwhich comprises two channels, one working on the in-phase signal and oneworking on the quadrature signal. Each channel has a mixer whichreceives the first radio frequency signal and a divide by 4 circuitwhich receives an output of the mixer and which itself provides an inputto a respective second input of the mixer. The output of the divide by 4divider is also provided to the respective in-phase and quadraturemixers. The feedback loop formed by the mixer and the divider by 4divider in fact forms a divide by 3 mixer, as is known to the personskilled in the art.

[0008] Preferably the GSM transmission path comprises a phase sensitivedetector having a first input for receiving an output of thein-phase/quadrature up-converter, and a second input for receiving anoutput of the image reject mixer in the phase locked loop. An output ofthe phase sensitive detector is provided to an input of a voltagecontrolled oscillator which in turn generates a radio frequency outputsignal. The output of the voltage controlled oscillator is provided to afurther input of the image reject mixer. Because the image reject mixerreceives both the in-phase and quadrature signals of the first signal,it can be electronically selected to output either the upper side bandor the lower side band. Thus the phase locked loop can be selectivelylocked to a frequency of F₁ minus the intermediate frequency or F₁ plusthe intermediate frequency.

[0009] The output of the voltage controlled oscillator in the GSM pathis also provided to a high power amplifier which can be driven in ClassC mode in order to obtain high efficiency. Driving the amplifier inClass C mode generates harmonics at multiples of the frequency of thevoltage controlled oscillator. However it is apparent that thesemultiples are spaced apart by at least 850 MHz and therefore can beeasily removed by relatively simple filtering.

[0010] The UMTS path may comprise one or more electronically controlledvariable gain amplifiers serving to amplify the modulated intermediatefrequency signal. The amplifiers typically only have to operate across arelatively narrow band of frequencies centered about the intermediatefrequency, e.g., 450 MHz. The amplifiers can therefore be constructed tobe particularly linear. The output from the amplifiers is then mixedwith the first signal by a single side band mixer in order to up-convertit to the UMTS output frequency in the range of 1.92 to 1.98 GHz. Theoutput from the mixer may then be passed through a further variable gainamplifier giving around 25 to 30 dB of gain. The output of the amplifiermay be passed through a surface acoustic wave filter before beingsupplied to a further off-chip power amplifier. A feedback path is alsoprovided for sampling the output of the power amplifier, detecting theoutput level thereof, digitizing it through an analog to digitalconverter and providing a measurement of the power output to a UMTSpower control logic circuit.

[0011] It is thus possible, in a preferred embodiment of the invention,to provide a multi-mode transmitter operable in both UMTS and GSMtransmission modes, the transmitter comprising:

[0012] a. a signal input for receiving a signal (such as an I/Q basebandsignal) to be upconverted by the transmitter;

[0013] b. an oscillator for generating a first radio frequency signalhaving a frequency F₁, where F₁ is 1.5 or 0.75 times the desired carrierfrequency F_(C);

[0014] c. a first divider for receiving the first radio frequency signalF₁ and dividing it by three so as to form an intermediate frequencylocal oscillator signal, I_(F), having a frequency substantially at 0.25or 0.5 times the desired carrier frequency;

[0015] d. at least one mixer for mixing the signal from the signal inputwith the intermediate frequency local oscillator signal to create amodulated intermediate frequency signal;

[0016] e. a GSM path responsive to the modulated intermediate frequencysignal; and

[0017] f. a UMTS path responsive to the modulated intermediate frequencysignal;

[0018] wherein the GSM path comprises a voltage controlled oscillatorwithin a phase locked loop, the phase locked loop including an imagereject mixer receiving at a first input thereof the output of thevoltage controlled oscillator, and at second inputs thereof in-phase andquadrature representations of the first radio frequency signal, suchthat it selectively outputs either the upper or lower side band as thesignal to the phase sensitive detector within the phase locked loop; and

[0019] wherein the UMTS path comprises at least one variable gainamplifier and a mixer for mixing the first radio frequency signal withthe modulated intermediate frequency signal and outputting one of theside bands to a further amplifier stage.

[0020] The oscillator may be followed by a divide by two stage such thatin-phase and quadrature versions of the divided oscillator signal can beeasily obtained. In such an implementation the oscillator frequencyneeds to be doubled to 1.5 or 3 times the desired carrier frequency.

[0021] According to a second aspect of the present invention, there isprovided a direct conversion multi-mode receiver comprisingelectronically reconfigurable filters.

[0022] It is thus possible, by providing reconfigurable filters, tomanipulate the output from the direct conversion multi-mode receiversuch that it is suitable for additional processing operations to beperformed on the signal in order to extract the data therein.

[0023] Advantageously the direct conversion multi-mode receiver alsocomprises offset generators which can be used to apply a controllableoffset to a summer which is responsive to the output of the receiver.

[0024] In the direct conversion topology, it is highly desirable inorder to be able to provide an offset to the output from the receiver. Areason for this is that the down conversion is performed by mixing thereceived radio signal, which is nominally centered about a frequencyF_(r), with a locally produced radio signal, also having a frequency ofF_(r). It therefore follows that the carrier of the received signal isdown converted to a DC signal (or very near DC in the case of oscillatorfrequency mismatch). Any spurious DC offset (or low frequency signal)therefore needs to be subtracted from the output of the converter inorder to reduce the dynamic range required of the subsequent processingcircuitry, which in reality will comprise analog to digital converterssuch that further processing can then be performed in the digitaldomain. The act of removing the DC offset means that the dynamic rangeand resolution of the converters required can be reduced, thereforereducing the cost of the subsequent processing circuitry.

[0025] In a preferred embodiment, a direct conversion multi-modereceiver is provided which comprises at least one electronicallyreconfigurable filter arranged to filter a base band signal receivedfrom the receiver, and an offset generator, each of the reconfigurablefilter and the offset generator is under the control of a controlcircuit such that the offset and filter response can be automaticallycontrolled as a function of reception mode and signal conditions. Thefilter may be implemented in hardware or software. Hardwareimplementations will tend to be preferred as they do not place so greata load upon post ADC processing resources.

[0026] According to a third aspect of the present invention, there isprovided a hybrid filter exhibiting substantially uniform group delay ina pass band thereof, wherein the filter comprises a combination of aChebychev and an inverse Chebychev response.

[0027] When designing an analog filter for communications applications,there are generally difficult trade-offs to be made between requirementsfor selectivity, group delay and complexity. Ideally, we want thephysical delay for a signal passing through the filter to be uniformirrespective of its frequency content. Thus, in terms of a phase versusfrequency graph, the phase delay needs to increase linearly withfrequency.

[0028] In the context of telecommunications, and in particular UMTScommunications, it is important that differential group delay be avoidedas this can give rise to inter-symbol interference.

[0029] However, it is also generally necessary to obtain goodselectivity, that is a rapid transition between the pass band and stopband.

[0030] It is well known that high Q filters that display goodselectivity performance such as Chebychev and Elliptic filters oftensuffer in terms of differential group delay performance. On the otherhand, filters that exhibit good differential group delay performancesuch as Butterworth and Bessel filters generally have inferior roll-offcharacteristics. The inventor has realized that combination of differentfilter characteristics can be arranged to give a desired response. Thenew filter design avoids the draw backs of conventional filtertechnologies and provides excellent selectivity, attenuation anddifferential group delay characteristics. The inventor has noted thatthe inverse Chebychev pass band characteristics are substantiallyidentical to those of the Butterworth, but the filter contains stop-bandzeros which give superior initial roll-off performance. On the otherhand, the Chebychev filter has good initial roll-off characteristics andvery high levels of ultimate attenuation.

[0031] Turning to the group delay characteristic, it is known to theperson skilled in the art that at frequencies well below the cut-offfrequencies, both the Chebychev and inverse Chebychev filters exhibitsubstantially uniform group delay. However, in the region of the cut-offfrequency the first derivative of the group delay for the two differentfilters has opposite signs. The inventor has realized that by combiningthe two characteristics it is possible to achieve substantialcancellation of the group delay characteristics in the region near thecut-off frequency. This cancellation can be arranged to be sufficient toensure that by the time the group delay cancellation begins to fail,thereby resulting in undesirable group delay characteristics, themagnitude of the signals have been sufficiently attenuated at thesefrequencies such that they become relatively unimportant.

[0032] The hybrid filter represents a good choice for a reconfigurableswitched band multi-mode filter whose modes of operation may placediffering demands on the filter performance. Thus, in the context of amulti-mode receiver, one mode of operation may place stringentrequirements on the filter in terms of differential group delay, but notsuch difficult requirements in terms of selectivity. The other mode ofoperation might have more stringent requirements in terms ofselectivity, but less difficult requirements in terms of differentialgroup delay. The hybrid Chebychev/inverse Chebychev filter proposed hereis an effective solution for both requirements. In addition to bandswitching the filter, it is also possible to adjust the relativecut-offs of the two constituent filters further optimizing performancein different modes of operation.

[0033] According to a fourth aspect of the present invention there isprovided a dual mode single chip transceiver comprising a transmitterand a receiver, wherein frequency synthesizers are shared by thetransmitter and the receiver, wherein the transmitter comprises anup-converter for receiving an input and up-converting it by mixing theinput with a first synthesized frequency, and wherein in a GSM mode anoffset phase locked loop is used to translate phase modulation at theup-converter output onto an RF carrier, and in a UMTS mode theup-converted signal is linearly amplified; and wherein the receivercomprises at least one direct down-converting channel for downconverting the received signal.

[0034] It is thus possible to significantly reduce the implementationcost of a dual mode GSM/UMTS transceiver architecture by sharing many ofthe transmit and receive components within an integrated circuit.

[0035] Preferably the UMTS transmitter includes a further frequencyup-conversion stage. Thus a first up-converter may produce an output atan intermediate frequency. The intermediate frequency may then belinearly amplified before being frequency up-converted to a final outputfrequency. This final output frequency may then be passed to a poweramplifier which is not integrated with the integrated circuit. Theprovision of one or more power amplifiers “off-chip” reduces the signalleakage between the transmitter path and the receiver paths. This isparticularly important as UMTS operates in full duplex mode andconsequently signal leak through could have a degrading effect onreceiver performance.

[0036] Preferably the UMTS duplex filter is also provided off-chip.

[0037] According to a fifth aspect of the present invention there isprovided an automatic gain controller for a multi-mode homodynereceiver, the controller comprising an open loop controller responsiveto a first signal for setting an initial gain, and a closed loopcontroller responsive to a measurement of signal power or amplitude formaintaining the signal power or amplitude at an output of the variablegain amplifier within a predetermined range.

[0038] It is thus possible to use an open loop controller to provide arapid initial set up of the amplifier in order to bring it to roughlyinto a desired operational state, and then to “fine tune” the gain ofthe amplifier using a feedback loop. This enhances the set up time ofthe amplifier when changing mode.

[0039] UMTS radio systems require continuous full duplex operation. Aspart of this, automatic gain control (AGC) is required to maintainacceptable performance under varying signal level and channelconditions. The use of a homodyne architecture places certainconstraints on the way in which the automatic gain controller canoperate.

[0040] Homodyne receivers are susceptible to generating unwanted DCoffsets in the in-phase/quadrature analog base band paths thereof. Inorder to maintain acceptable performance it is necessary to remove theseDC offsets as they are indistinguishable from an on-channel signal.Under weak signal conditions, it is quite normal for the DC offset to besubstantially larger in magnitude than the wanted signal. Thus, if theDC offsets were not removed, acceptable reception of the wanted signalis likely to prove impossible. A common approach to removing suchoffsets in wide band receivers is to use simple high pass filters or ACcoupling (DC blocking) circuit arrangements.

[0041] Automatic gain control is achieved in homodyne receivers byadjusting the gain of analog base band amplifiers. However, adjustingthe gain also adjusts the DC offset levels and thus creates a transientsettling time problem with high pass filters which were introduced inorder to block the DC component. As a consequence, a low band-widthautomatic gain control loop is desirable in order to minimize transienteffects as these would occur as a result of shifts in the DC offset.However a low band-width automatic gain control is incompatible with therequirement for initial rapid acquisition of the signal.

[0042] The inventor has realized that it is possible to overcome thiswith a combination of a “feed-forward” or open loop automatic gaincontroller and a feedback automatic gain controller.

[0043] The feed-forward/open loop controller can either set up itsinitial values based on an initial measurement of the signal power madeby a suitable measuring device, such as a full wave or half waverectifier located at a suitable place in the receiver, for example atthe output of channel select filters, or alternatively the open loopcontroller may make an initial gain control setting based on the desiredmode of operation of the receiver. The open loop controller onlyoperates once, at each mode change, to set up the initial parameters ofthe variable gain amplifiers and other components which may require gainset up. From then on, control is passed to the closed loop which makesfine adjustments to the various gain levels.

[0044] According to a further aspect of the present invention, there isprovided a homodyne receiver comprising a high pass filter having avariable time constant and a variable gain amplifier, the filter andamplifier being upstream of an analog to digital converter, wherein whena step change of the gain of the variable gain amplifier is implementedthe time constant of the high pass filter is reduced for a predeterminedtime period.

[0045] According to a further aspect of the present invention, there isprovided a homodyne receiver comprising at least one signal conditionerupstream of an analog to digital converter and a high pass filter havinga variable time constant, wherein when the at least one signalconditioner is operated to cause a variation to be made to the signalsupplied to the analog to digital converter, the time constant of thefilter is set to a reduced value.

[0046] The highpass filter may be implemented downstream of the analogto digital converter and may therefore be implemented in the digitaldomain.

[0047] In a homodyne direct conversion receiver, it is necessary to usea high pass filter to remove DC offsets from the analog base band signalpaths. For a 3.84 MHz UMTS signal, the cut-off frequency for this highpass filter should be in the order of 10 kHz or so in order to removethe minimum level of the wanted signal energy. However, such a lowcut-off frequency implies a long settling time for DC transients. Theinventor has realized that the settling-time for DC transients can bemuch improved if the cut-off frequency of the filter is temporarilyincreased. This significantly enhances the receiver settling time.

[0048] The present invention will further be described, by way ofexample, with reference to the accompanying drawings, in which:

[0049]FIG. 1 is a circuit diagram schematically illustrating a combinedGSM and UMTS transmitter, with shared RF components integrated into asingle circuit;

[0050]FIG. 2 is a schematic diagram of the direct conversion multi-modereceiver;

[0051]FIG. 3 is a graph comparing the group delays of low pass Chebychevand inverse Chebychev filters versus frequency;

[0052]FIG. 4 is a graph of the magnitude response of low pass Chebychevand inverse Chebychev filters versus frequency;

[0053]FIG. 5 shows the combined magnitude response of the filters ofFIG. 4;

[0054]FIG. 6 schematically illustrates a combined GSM/UMTS transceiver;and

[0055]FIG. 7 schematically illustrates an automatic gain control.

[0056]FIG. 1 schematically illustrates transmitter circuitry accordingto an embodiment of the first aspect of the present invention. Thetransmitter can be selectively operated as either a dual band GSMtransmitter, or alternatively as a 3 G UMTS transmitter. The transmittercomprises a single chip 1 on which is located a single radio frequencysynthesizer 3. In accordance with an input signal f_(in) received by thesynthesizer 3, the synthesizer controls a local oscillator 5 such thatthe oscillator 5 produces a substantially fixed frequency signal that isat either 1½ or 3 times the desired carrier frequency, depending on theactual desired operating frequency for the GSM or UMTS transmissionpath. It should be emphasized that the actual frequency of the signalgenerated by the general oscillator 5 does not substantially differdepending upon the band of transmission, only its ratio to the desiredcarrier frequency. The frequency of the signal generated by the localoscillator 5 may, for example, be approximately 2.7 GHz and morespecifically in the range of 2.565 to 2.970 GHz. This corresponds to amodest adjustment of ±10% (in fact around 7% in this example) of thefrequency generated by the local oscillator.

[0057] The signal from the local oscillator 5 is then fed to a frequencydivider 7 that divides the frequency of the signal by 2. The frequencydivider 7 also allows the signal to be split into in-phase (I) andquadrature (Q) components. The I and Q components are fed into a furtherfrequency divider 9 in the form of a regenerative divider that isarranged to divide the input frequency by 3. The divide by 3 frequencydivider 9 comprises a frequency mixer 11 that receives the I and Qcomponents of the input frequency signal and mixes them withcorresponding components of the same signal that have been furtherfrequency divided by 4 by a further frequency divider 13. Thearrangement of the mixer 11 and divide by 4 divider 13 produces adivided by 3 frequency signal of the original input signal at the outputof the divide by 4 divider 13. The I and Q components output from thedivide by 3 divider 9 are therefore at ¼ or ½ of the carrier frequency.Taking our example of the input frequency f_(in) of 2.7 GHz, the I and Qcomponents at the output of the divide 3 divider 9 are at 450 MHz.

[0058] The individual I and Q components are supplied to respectiveindividual mixers 15 and 17 that each also receive the respective I andQ components of the analogue source signal. The mixers 15 and 17 act tocombine the respective source signal components together with the I andQ components at 450 MHz to generate a 450 MHz modulated intermediatefrequency signal which it then passed through a bandpass filter 19.

[0059] The intermediate frequency at the output of the bandpass filter19 is then propagated along two individual signal paths, each path beingused to generate respectively the GSM signal or the 3 G UMTS signal.

[0060] For GSM, the 450 MHz intermediate frequency signal is fed to afirst input of a phase comparator 21. The second input of the phasecomparator 21 also receives a 450 MHz signal that is received from abandpass filter 23. The input to the bandpass filter 23 is derived froman image reject mixer 25. The image reject mixer 25 receives as firstinputs the I and Q components of the 1.35 GHz signal that is obtainedfrom the combination of synthesizer 3, local oscillator 5 and divide by2 divider 7. The other input to the image reject mixer 25 is the carrierfrequency signal that is generated by a voltage controlled oscillator 27that is in turn controlled by the output from the phase comparator 21.The phase comparator 21, voltage controlled oscillator 27 and imagereject mixer constitute a phase locked loop that is used to translatethe modulated intermediate frequency to the radio frequency carriersignal. The phase locked loop acts as a tracking bandpass filter andthus removes the need for an RF bandpass filter such as a SAW or ceramicfilter that would otherwise be required to minimize out of bandemissions.

[0061] The carrier frequency is substantially either 900 MHz or 1.8 GHz,depending upon the GSM band on which it is desired to transmit. Theimage reject mixer can selectively either subtract the 900 MHz signalfrom the 1.35 GHz I and Q component signals to arrive at the 450 MHzsignal input to the bandpass filters 23, or to subtract the 1.35 GHzfrom the 1.8 GHz carrier signal, thus also arriving at a 450 MHz output.Thus the two signals input to the phase comparator 21 are always at 450MHz, i.e., the intermediate frequency. It is thus possible to cover theGSM 850/900 range and the GSM 1800/1900 range.

[0062] The modulated GSM signal is applied to a high power amplifier 29that receives a power control signal on an input line 31. The output ofthe amplifier 29 is fed to an antenna 33 via a switch 35.

[0063] For 3 G UMTS transmission, the intermediate frequency signaloutput from the bandpass filter 19 is fed through a pair of seriallyconnected variable gain amplifiers 37, 39 prior to being mixed with the1.35 GHz signal derived from the local oscillator 5 and divide by 2divider 7. The two signals are mixed at a single side band mixer 41 toproduce a 1.8 GHz signal that is itself further amplified using afurther single variable gain amplifier 43. Amplifiers 37, 39 and 43 arecontrolled using a power control circuit 45 that receives a sampled anddigitized input signal representative of the power of the transmittedoutput signal obtained via a tap at the output of amplifier 49. The 1.8GHz signal from the power amplifier 43 is passed through a UMTS RF SAWfilter 47 and a further UMTS power amplifier 49 before being transmittedvia the antenna 33. The signal is also passed through a duplexer andisolator unit 51 connected between the antenna and the UMTS poweramplifier 49 which selectively allows a received signal at the antennato be directed towards a receiver circuit.

[0064] The great advantage of the circuit described above is that asingle RF synthesizer, running at 2.7 GHz in the above example, is allthat is required for the generation and transmission of both a dual mode(850/900 and 1800/1900) GSM signal and a 3 G UMTS signal. The use of thedividers and mixers ensures that the desired carrier and outputfrequencies are always at fixed multiples of the synthesizer frequency.Thus is possible to provide a combined dual mode GSM and 3 G UMTStransmitter in a single circuit that has a relatively large number ofcommon circuit components for both transmission paths.

[0065]FIG. 2 schematically illustrates a direct conversion (also knownas homodyne) multi mode receiver. The receiver comprises two channels,generally indicated 100 and 102 for convenience. Schematically, thechannels are identical so only the first channel will be described indetail for convenience. However, in terms of operating performance, thechannels may operate at different frequencies, for example around 800 to1000 MHz for GSM 850 and GSM 900 and around 1.7 to 2.2 GHz for GSM 1800,GSM 1900 and UMTS. In these circumstances the individual components ofthe channels may be tailored in order to operate at their respectivebands. Each channel comprises a band pass filter 110 and 100 a whichserves to reject signals outside the pass band of the receiver. Thus,the filter 110 for the first channel 100 may be centered 900 MHz,whereas the filter 110 a for the second channel 102 may be centeredaround 1.8 GHz or so. These filters are necessary to stop powerful outof band transmissions from driving the receiver into saturation. Theoutput of the band pass filter 110 is provided to an input of anamplifier 112 whose output is provided to the first inputs of mixers 114and 116, respectfully. Second inputs of the mixers 114 and 116 receivein-phase and quadrature versions of a locally generated carrier signal.The in-phase and quadrature versions of the signal are generated by aphase shifter 118 which itself receives the locally generated signalfrom the combination of a multi mode fractional synthesizer 120, avoltage controlled oscillator 122 and a multi mode fractional frequencymultiplier 124. The components 120, 122 and 124 are shared by each ofthe channels 100 and 102. The mixers 114 and 116 mix the locallygenerated reference signal together with the received radio signal inorder to form the difference frequency therebetween. Since both theradio frequency and the locally generated frequency are nominally at thesame frequency, the information in the radio frequency is directly downconverted to base band. The base band signal is provided at the outputof each of the mixers 114 and 116. The outputs of the in-phase mixers114 and 114 a are provided to a first input of an in-phase summer 130which is shared between both channels. Similarly, the outputs of thequadrature mixers 116 and 116 a are provided to the first input of aquadrature summer 132 which is also shared between both channels. Eachof the summers 130 and 132 is also connected to receive an offset signalgenerated by a respective digital to analog converter 134 and 136. Theability to provide an offset is important since local oscillatorcoupling to the RF inputs could generate a DC offset, the size of whichdepends on both the magnitude and phase of the spuriously coupled signalwith respect to the locally generated reference. The output of thesummer 130 is provided to the input of an electronically controlled lowpass filter 140 which in turn is followed by an electronicallycontrolled variable gain amplifier 142. Similarly the output of thesummer 132 is provided to an electronically controlled low pass filter150 and a variable gain amplifier 152. The outputs of the amplifiers 142and 152 are provided to respective analog to digital converters 160 and162 whose digital outputs, after filtering in respective finite impulseresponse filters are provided to a signal processing and control unit170. For each of the in-phase the quadrature channels, the control unitestimates the offset that is required for each of the channels andprovides an offset signal to the digital to analog converters 134 and136. The control unit 170 also sets up the filter characteristics of theswitched filters 140 and 150 in the way appropriate to the operatingmode of the receiver.

[0066] It is thus possible to provide a multi mode direct conversionreceiver architecture as shown in FIG. 2. The architecture has theadvantage that much of the receiver hardware is reused for differentmodes of operation. The reuse of common hardware can provide asignificant cost and power saving when compared with duplicatingfunctionality as has hitherto been the case.

[0067] The analog base band sections are designed so that they can bereconfigured to meet the requirements of the various transmissionsystems. This means that the architecture must allow for the:

[0068] i. reconfiguring of the gain line up,

[0069] ii. reconfiguring of the channel filters

[0070] iii. reconfiguring of the analog to digital converters speed andresolution, and

[0071] iv. reconfiguring of the DC offset compensation.

[0072] As noted hereinbefore separate radio frequency low noiseamplifiers and in-phase/quadrature down converters are utilized as theseneed to be optimized for the frequency and mode of operation. Althoughin principle it is practical to have a reconfigurable RF front end fordifferent modes of operation, it is believed to be more cost effectiveat the present time to have dedicated RF front ends.

[0073] As noted hereinbefore, it is highly desirable that the analogprocessing circuitry within a mobile telephone has filters includedtherein which can give both rapid attenuation between the pass band andthe stop band, and which also exhibit good group delay, and specificallydo not exhibit differential group delay so as to avoid inter-symbolinterference. FIG. 3 compares the group delays of an inverse Chebychevfilter, represented by line 200, and the group delay of a Chebychevfilter, represented by line 202, plotted as a function of frequency.Frequency units have been included on the ordinate of FIGS. 3, 4 and 5such that the various graphs can be easily compared. It will be seenthat, in the example given in FIG. 3, the inverse Chebychev group delaystarts to increase for frequencies in excess of 2×10⁵ radians persecond. However, the Chebychev group delay starts to decrease forangular frequencies in excess of 2×10⁵ radians per second and thisdecrease continues until the frequency 10⁶ radians per second, then theChebychev group delay increases sharply. It will be appreciated that inthe limited region 204 where the group delays are of an opposite sign,partial cancellation of the group delay characteristics can be achievedthereby effectively extending the region 206 extending from lowfrequencies up to approximately 2×10⁵ radians per second where the groupdelays are essentially invariant with respect to frequency to anincreased upper frequency of 10⁶ radians per second.

[0074]FIG. 4 schematically illustrates the magnitude response for theinverse Chebychev and Chebychev filters whose group delays were shown inFIG. 3. It can be seen that the inverse Chebychev filter in this examplehas a substantially flat magnitude response up to approximately 10⁶radians per second, and then the magnitude response falls steeplytowards a notch occurring at 5×10⁶ radians per second. At this point,the magnitude of the response is suppressed by over 60 dB. The Chebychevfilter also has a substantially uniform magnitude response in the passband, but has a roll over frequency of approximately 2×10⁶ radians persecond from where the magnitude falls off rapidly.

[0075]FIG. 5 shows the combined magnitude response of aChebychev/inverse Chebychev filter using the individual filters shown inFIGS. 3 and 4. It is seen that the combined response 210 issubstantially flat up to 10⁶ radians per second and then falls steeply,being approximately 10 dB down by 2×10⁶ radians per second and over 60dB down by 5×10⁶ radians per second. Furthermore, the group delay can bemaintained as substantially constant up to 1×10⁶ radians per second.

[0076] It is appreciated by the person skilled in the art that analogfilter design is an immensely complex mathematical exercise. It is,however, also well known that many filter designs have already beenanalyzed and described in a normalized form such that an engineer caneffectively use a “recipe” of a standard form to design a specificfilter characteristic. Furthermore, computer aided design packages nowalso allow for filter characteristics to be accurately depicted. Forthese reasons, the specifics of the design do not need to be describedin detail as sufficient support exists in the prior art to enable theperson skilled in the art to implement the filter. However, for thespecific filters whose responses are shown in FIGS. 3 to 5, the Polepositions for Inverse Chebychev response were calculated by firstcalculating the pole positions for Chebychev response and using polereciprocation. $p = \begin{bmatrix}{1.204 + \quad {2.258i}} \\{2.408\quad} \\{\quad {1.204 - \quad {2.258i}}} \\{{{\,^{-}1.204} - \quad {2.258i}}\quad} \\{{\,^{-}2.408}\quad} \\{{{\,^{-}1.204} + \quad {2.258i}}\quad}\end{bmatrix}$

[0077] Valid poles exist in the negative half of the S-plane$z = {{\begin{bmatrix}{0} \\{1.155i} \\0 \\{1.633 \cdot 10^{16_{i}}}\end{bmatrix}\quad \omega \quad z\text{:}} = {z_{1}}}$

[0078] Position of stopband zero

[0079] The Inverse Chebychev Transfer function used was${{Hic}_{1}\text{:}} = \frac{\lbrack {( \frac{1}{p_{3}} ) \cdot ( \frac{1}{p_{4}} ) \cdot \frac{1}{p_{5}}} \rbrack \cdot \lbrack {( \frac{s_{i}}{\omega c} )^{2} - ( z_{1} )^{2}} \rbrack \cdot \lbrack {( \frac{s_{i}}{\omega c} )^{2} - ( z_{3} )^{2}} \rbrack}{\lbrack \lbrack {( {\frac{s_{i}}{\omega c} - \frac{1}{p_{3}}} ) \cdot \lbrack {( \frac{s_{i}}{\omega c} ) - ( \frac{1}{p_{4}} )} \rbrack \cdot \lbrack {( \frac{s_{i}}{\omega c} ) - ( \frac{1}{p_{5}} )} \rbrack \cdot \lbrack ( z_{1} )^{2} \rbrack \cdot ( z_{3} )^{2}} \rbrack \rbrack}$

[0080] The Pole Positions for solution to Chebychev Polynomial were${p1} = \begin{bmatrix}{0.313 + \quad {1.022i}} \\{\quad 0.626\quad} \\{\quad {0.313 - \quad {1.022i}}\quad} \\{{{\,^{-}0.313} - \quad {1.022i}}\quad} \\{{\,^{-}0.626}\quad} \\{{{\,^{-}0.313} + \quad {1.022i}}\quad}\end{bmatrix}$

[0081] Valid poles exist in the negative half of the S-plane

[0082] The Chebychev Transfer Function used was${{Hc}_{i}\text{:}} = \frac{{p1}_{3} \cdot {p1}_{4} \cdot {p1}_{5}}{\lbrack {{( {\frac{s_{i}}{\omega c1} - {p1}_{3}} ) \cdot ( {\frac{s_{i}}{\omega c1} - {p1}_{4}} )}( {\frac{s_{i}}{\omega c1} - {p1}_{5}} )} \rbrack}$

[0083] The Transfer Function of Hybrid Filter is the combination of thetwo individual transfer functions:

X _(i):=(Hc _(i) ·Hic _(i))

[0084] Being a hybrid filter, each of the two elements of the filter canbe individually adjusted to better tailor the response of the filter tomeet the needs of the particular application

[0085] For example, with the Chebychev response, it is possible toadjust the following:

[0086] 1. The cut-off frequency

[0087] 2. The filter order (number of poles)

[0088] 3. The in-band ripple

[0089] For the Inverse Chebychev filter, it is possible to adjust thefollowing:

[0090] 1. The minimum stop-band attenuation

[0091] 2. The maximum tolerable pass-band roll off

[0092] 3. The relative frequency at which the minimum stop-bandattenuation is reached

[0093] 4. The filter order

[0094] The filter order can be adjusted by implementing the filter as acascade of filter stages (i.e., the stages are connected one to another)whereby one or more stages can be switched out of the cascade such thatthey are bypassed. Thus bypassing a stage reduces the order of thefilter.

[0095]FIG. 6 schematically illustrates a dual mode GSM/UMTS transceiverconstituting an embodiment of the present invention. The transceivergenerally comprises a transmission channel 300 and a reception channel302. Although represented in simplified form in FIG. 6, the transmissionchannel 300 actually contains the dual mode transmitter shown in FIG. 1of the present drawings. In order to simplify the understanding of FIG.6, those parts of FIG. 6 which are similar to parts shown in FIG. 1 willbe given like reference numerals thus, it can be seen that thetransmitter receives a frequency synthesized signal from a synthesizer3, that this synthesized signal is passed through a divide by threefrequency divider 9 before being provided to the in-phase and quadraturemixers 15 and 17 of the up-converter. The output of the up-converter is,in a GSM mode supplied to a phase locked loop frequency shiftercomprising a phase detector 21, a voltage controlled oscillator 27, amixer 25 and a band pass filter 23. Thus the operation of thesecomponents is as described hereinbefore with reference to FIG. 1. It maybe noticed that there are further divide by two frequency dividers 306,308 and 310 shown in the transmitter schematic, but it will beappreciated that these have little overall effect on the final operationof the transmitter, and in particular that frequency dividers 308 and310 effectively nullify each other, although they do allow for the markspace ratio of the wave forms to be converted to an ideal 50-50.Similarly, the UMTS path comprises an array of linear amplifiers and afrequency up-converter. The components labeled 37, 39, 41 and 43 asshown in FIG. 1 are schematically represented by the box 320 in FIG. 6.

[0096] It will be appreciated that the receiver section 302 is asimplified representation of FIG. 2. Here the multi-mode fractionalfrequency multiplier 124 of FIG. 2 is also embodied within the dualfraction synthesizer 3 shown in FIG. 6. It can also clearly be seen thattwo channels are provided, each having an in-phase and quadrature mixerfor frequency down-converting the received signal to the base band. Thecomponents 130, 132, 134, 136, 140, 142, 150, 152, 160 and 162 of FIG. 2are schematically represented by the box 330 in FIG. 6. It is clear thatboth the transmitter channels share components, both the receiverchannels share components, and indeed that the receiver and transmittersshare the frequency synthesizer components. This sharing of componentsallows for cost reductions in the final price of the integrated circuit,and also reduces the overall power consumption of the transceivercompared to implementations where each of these components are providedfor their individual functions, and hence effectively duplicated.

[0097] GSM works on a time division duplex system and hence thetransmitter and receiver are not operating concurrently. However UMTSoperates on full duplex and consequently the transmitter and receiver dooperate concurrently. In the UMTS mode it is important to limit thedesensitization of the receiver due to interactions with the UMTStransmitter. This is achieved firstly by not integrating the low noisepower amplifier with the chip itself, and secondly by limiting thetransmitter power at the transmitter output pins. Furthermore, it isimportant to ensure that the noise level in receive band at the transmitoutput is adequate. For example, with an external low noise amplifierhaving 13 dB of gain, the transmit power being limited to +3 dBm at thetransmitter output pin and a transmit noise floor in the receive band of−140 dBc/Hz if isolation of 30 dB is obtained between the transmitterand receiver then the impact of the transmitter on receiver sensitivitywill be approximately 0.1 dB.

[0098] The architecture shown in FIG. 6 is operable in several modes,the frequency plan for the transceiver may be as follows:

[0099] GSM 850/900 receive mode—the synthesizer frequency is three timesthe RF frequency.

[0100] GSM 1800/1900 receive mode—the synthesizer frequency is 1.5 timesthe carrier frequency.

[0101] UMTS receive mode—the synthesizer frequency is 1.5 times the RFcarrier frequency.

[0102] GSM 850/900 transmit mode—the synthesizer frequency is 3 timesthe RF carrier frequency and the intermediate frequency is a ½ of the RFcarrier frequency.

[0103] GSM 1800/1900 transmit mode—the synthesizer frequency is 1.5times the RF carrier frequency and the intermediate frequency is ¼ ofthe RF carrier frequency.

[0104] UMTS transmit mode—the synthesizer frequency is 1.5 times the RFcarrier frequency and the intermediate frequency is ¼ of the RF carrierfrequency.

[0105]FIG. 7 schematically illustrates a hybrid feed forward and feedback automatic gain control system constituting an embodiment of thepresent invention. The circuit diagram shows a variable gain amplifierwhose output is supplied to the hybrid Chebychev/inverse Chebychevfilter, generally indicated 402, which has hereinbefore been described.An output of the hybrid filter 402 is provided to the input of a furthervariable gain amplifier which is schematically represented as threeindependently electronically controllable variable gain amplifiers 404,406 and 408 which together serve to provide a variable gain between zeroand 54 dB in one dB steps. An output of the final amplifier 408 isprovided as an output 410 from the automatic gain controller.

[0106] The feed forward controller, generally indicated 420, comprises areceived signal strength indicator (RSSI) log strip which has an inputconnected to the output of the hybrid filter 402. The RSSI log strip isused to estimate the signal strength at the output of the hybrid filter.The RSSI log strip 422 produces as an output thereof 424 a voltage whichis substantially linearly proportional to the composite power of thesignal at the output of the filter expressed in dBm. This output signalis filtered by a low pass filter 426 before being supplied to the analoginput 428 of a six bit analog to digital converter 430. The signal atthe input 428 of the analog to digital converter 430 is digitized inresponse to a “start convert” signal and the output of the conversion issupplied to a filter gain logic controller 432. The filter gain logiccontroller has two outputs, one of which is supplied to a register 434for controlling the gain of the variable gain amplifier 400, whilst theother output is supplied as a 6 bit word to a 6 bit updown counter 440.Thus, the counter can be loaded with the output of the filter gain logiccontroller 432.

[0107] The signal in a UMTS receiver is a composite of the wantedsignal, noise and any residual interfering signals. This compositesignal appearing at the output of the hybrid filter 402 is digitized toproduce a digital word that has the characteristic of “A” dB per bit,where A represents an arbitrary number. This control word is used to setthe gain of the variable gain amplifiers. The variable gain amplifiersare designed to have a gain reduction characteristic of “A” dB per bit.As such, if the signal level into the receiver increases by 5 A dB, thenthe digitized control word will also increase by 5. This in turn willresult in the reduction of the gain of the variable gain amplifiers 404,406 and 408 by a composite gain amounting to 5A dB. Thus the signallevel at the output of the variable gain controller 410 is keptsubstantially constant. Because the approach described so far uses feedforward techniques, there are no band-width implications that mightaffect a feed back system. Thus, it is possible to rapidly set up gainfor the automatic gain controller for example when switching modes ofoperation or channels. Having rapidly acquired the composite signallevel and setting up an initial gain, further gain control is performedby a feedback scheme.

[0108] The feedback controller, generally indicated 450, comprises arectifier 452 connected to the output of amplifier 408 in order todeduce a rectified signal representing the signal power at the output ofthe amplifier 408. The signal from the rectifier 452 is low passfiltered by a filter 454 before being supplied to the inputs of a windowcomparator 456 which, as recognized by the person skilled in the art,compares the signal at the input therein with high and low thresholdsdefining a window and produces an output 458 which is indicative ofwhether the signal is below the window threshold or above the windowthreshold. The signal 458 is provided to a count direction control input(up/down input) of the counter 440. As shown in FIG. 7, the windowcomparator 456 is also configured to provide an output which indicateswhen the input thereto is within the bounds defining the window, andthis output is sent to an AND gate 460 which serves to gate theprovision of a clock signal to the counter 440. Thus, when the output410 is within the power band defined by the window comparator 456, thecounter 440 is inhibited from receiving its clock signal. The clocksignal 462 is also ANDed with an “enable feedback” signal at an AND gate464 which gives overall control of whether the feedback loop shouldoperate or not. The output of the AND gate 464 is provided to an inputof the AND gate 460, the output which is connected to the clock pin ofthe counter 440. The output of the counter 440 is provided to a gaindecoder 470 which in turn sets the gains of the amplifiers 404, 406, and408.

[0109] In use, the signal level detector and window comparator is usedto ensure that the composite signal level and the amplifier outputs iskept within a narrow range, for example +/−0.5 dB. If the compositesignal level is above the threshold of the window comparator, theup/down counter is enabled and the gain is adjusted on every clockcycle. As such, the clock sets the time constant of the feedback loop.If the composite signal level is below the window comparator threshold,then the up/down counter is also enabled, but this time counts in theopposite direction. As a consequence, the feedback loop will act toalways ensure that the composite signal level is tightly controlled andadjusted at a rate determined by the frequency of the clock 462.

[0110] Referring to FIG. 2, it should be noted that every time a gaincontrol adjustment is made or a offset correction is made, this stepchange can result in a transient DC offset appearing at the output ofthe filter. The DC offsets could be removed by a high pass filter, butfor a truly generic receiver it is better that the DC offset correctionshould be performed in the digital domain. Thus at each gain change anew DC offset is estimated and the estimate is supplied to theconverters 134 and 136 such that a correction is added at the summers130 and 132. This said, the transient occurring as a result of gainchange still has an unwanted effect of the low pass filters 140 and 150.This transient decays in time, but during the settling time of thefilter the homodyne receiver is effectively blinded. This is because theconversion range of the analog to digital converter will necessarily belimited, and the offset may cause the converter to have to convertoutside its nominal operating range.

[0111] Given that the receiver is effectively non-functional during thistransient period, the inventor has realized that it is permissible tochange the filter characteristics during the short period in order toallow for a more rapid settling time. Thus, when a change in gain oroffset is implemented, the high pass filter is simultaneously, or nearsimultaneously set to a wide bandwidth such that the DC transient willquickly settle. The settling time can be estimated from knowncharacteristics of the filter. After the settling time, the filter isautomatically switched back to the nominal required setting for itsproper operation. The filter can be under to the control of a timer, forexample implemented as a monostable, which alters the filtercharacteristics for a brief but well defined period. This techniqueensures the fastest possible receiver settling time whilst at the sametime minimizing the amount of wanted modulation energy that might beremoved by the high pass filter. A typical ratio between the twobandwidth settings might be in the order of 10 to 1, although this isonly a non-limiting example and other ratios may be chosen by thedesign. The implementation of this technique is not dependent on thefilter technology. Thus the filter might be implemented as a switchedcapacitor filter, a switched bandwidth active R-C filter,gyrator-capacitor filter and so on. The specific implementation of thefilter is within the knowledge of the person skilled in the art.

[0112] It is thus possible to provide a multi mode receiver andtransceiver which are particularly suited for use in mobile telephony.

1. An automatic gain control for a UMTS homodyne receiver, comprising anopen loop controller responsive to a first signal for setting an initialgain, and a closed loop controller responsive to a measurement of signalpower or signal amplitude for maintaining the signal power or amplitudeat an output of a variable gain amplifier within a predetermined range.2. An automatic gain control as claimed in claim 1, in which the openloop controller sets a gain of a first variable gain amplifier.
 3. Anautomatic gain control as claimed in claim 1, in which the open loopcontroller sets up an initial gain each time a mode change occurs.
 4. Anautomatic gain control as claimed in claim 3, in which the open loopcontroller sets up an initial gain in response to a measurement ofsignal strength.
 5. An automatic gain control as claimed in claim 3, inwhich the open loop controller set up an initial gain based on the modeof operation of the receiver.
 6. An automatic gain control as claimed inclaim 1, in the closed loop controller includes a window comparatordefining an acceptable signal power or signal amplitude, and the closedloop acts to vary the gain of a variable gain amplifier when the signalpower or amplitude is outside the range defined by the windowcomparator.
 7. An automatic gain control as claimed in claim 6, in whichthe gain controller includes a counter which holds a value forcontrolling the gain of the variable gain amplifier, and in which aclock signal is supplied to the counter or the counter enabled only whenthe window comparator determines that the signal power or amplitude isoutside of the range defined by the window comparator.
 8. An automaticgain control as claimed in claim 1, in which the closed loop isimplemented in digital form and the operation of the closed loop can beenabled or disabled by a control signal.
 9. An automatic gain control asclaimed in claim 1, wherein the variable gain amplifier is formed from aplurality of variable gain amplifiers in series.
 10. A direct conversionreceiver including an automatic gain control as claimed in claim
 1. 11.A mobile telephone operable in GSM and UMTS modes, including anautomatic gain control as claimed in claim
 1. 12. An automatic gaincontrol for a multi-mode receiver, the gain control comprising a feedforward controller which sets up an initial gain based on a measurementof signal strength or upon mode information, and a feedback controllerwhich then has responsibility for maintaining a signal at a desiredamplitude.
 13. An automatic gain control as claimed in claim 12, inwhich the feedback controller includes a window comparator and acts tovary the gain of a variable gain device when the signal amplitude isoutside an acceptable range defined by the window comparator.
 14. Anautomatic gain control as claimed in claim 12, in which the feedbackcontroller is implemented in the analog domain and comprises at leastone of a proportional, integral and differential control function.